Current sensor including an integrated circuit die including a first and second coil

ABSTRACT

A current sensor includes a coils located within the integrated circuit die and inductively coupled to a conductor located in the integrated circuit package holding the die. The inductors sense the current in the conductor and supply the sensed signal to an integrator that supplies a voltage indicative of the current in the conductor.

This application is a continuation of U.S. patent application Ser. No.11/428,082, now U.S. Pat. No. 7,397,234, entitled “CURRENT SENSOR WITHRESET CIRCUIT,” filed Jun. 30, 2006, naming inventors Donald E. Alfanoand Timothy J. Dupuis, which application is hereby incorporated hereinby reference.

This application also relates to U.S. patent application Ser. No.11/311,603, entitled “INTEGRATED CURRENT SENSOR,” filed Dec. 19, 2005,naming inventors Donald E. Alfano and Timothy J. Dupuis; and to U.S.patent application Ser. No. 11/311,517, entitled “INTEGRATED CURRENTSENSOR PACKAGE,” naming inventors Timothy J. Dupuis and John Pavelka,filed Dec. 19, 2005, which applications are incorporated herein byreference.

BACKGROUND

1. Field of the Invention

The present invention relates to current sensors, and more particularly,to a current sensor having an improved reset circuit.

2. Description of the Related Art

Within various circuit implementations, such as power supplies, there isoften a need to detect a current provided at a particular point within acircuit. For example, a detected current may be used as feedback forcontrolling other parts of a circuit. Various techniques are presentlyused to sense currents within electronic circuits, but each of thesetechniques have shortcomings. One approach, illustrated in FIG. 1,utilizes a resistor 102 connected across the inputs of an operationalamplifier 104 to provide a voltage V_(SENSE) that may be used todetermine a current 106. A low value resistor, in the range of 10 mOhms,may be implemented. However, a drawback of this approach is the highloss provided by the circuit. The high loss may be mitigated by reducingthe value of the resistor 102, however, this also reduces the signalV_(SENSE) that may be detected. While this type of circuit may be usedto sense current in direct current (DC) applications, the resistor 102has usually not been capable of being readily integrated.

Referring now to FIG. 2, a further prior art system, utilizing a Halleffect device 202, connected across the inputs of an operationalamplifier 204, is illustrated. The Hall effect device 202 generates avoltage across the inputs of the operational amplifier 204, responsiveto a current 206, to provide an output signal V_(SENSE). While thisapproach has a relatively low loss and may be used to detect directcurrent (DC), the use of the Hall effect device 202 generally provides acircuit having a higher cost. Furthermore, accuracy and noise issues aregenerally greater in current sensors that implement a Hall device, asthe Hall voltage is a relatively small value.

With reference to FIG. 3, a current sensor that uses a magneto resistivesensor is illustrated. The magneto resistive sensor consists of amagneto resistive element 302 connected across the inputs of operationalamplifier 304 to detect a current 306. The magneto resistive element 302has the property that the resistance of the element changes with respectto the magnetic field caused by the current 306. This circuit requiresthe use of special technology which raises the cost of the device.Additionally, accuracy issues arise even though the current may besensed with very low loss.

Referring now to FIG. 4, an alternative prior art technique fordetecting current, through the use of a current transformer 402, isillustrated. As is shown, the current transformer 402 has a primary side404 with a single loop and a secondary side 406 with multiple loops. Aload resistance 408 is in parallel with the secondary side 406 of thetransformer 402. The current transformer 402 is used to detect a current410. The transformer 402 creates an output current equal to I_(p)/n,with I_(p) being the detected current and n being the turns ratio of thetransformer 402. In this configuration, the resistance of the secondaryside of the transformer is reflected to the primary side with the ratio1/n². While current transformers work well for detecting currents, theyare large and have a medium loss level and only work with alternatingcurrent (AC) circuits.

Another method for measuring currents involves the use of a Rogowskicoil. Unfortunately, the voltage induced in a Rogowski coil is verysmall and easily disturbed when a measured current is less than, forexample, 100 Amps. However, a Rogowski current transducer has a numberof advantages over the current transformer illustrated in FIG. 4. Forexample, the Rogowski current sensor is linear, has no core saturationeffects, has a wide bandwidth and a wide measurement range and is arelatively simple structure. The Rogowski coil comprises a toroidalwinding placed around a conductor being measured. The Rogowski coil iseffectively a mutual inductor coupled to the inductor being measured,where the output from the winding is an EMF proportional to the rate ofchange of current. While the above described techniques provide anindication of a sensed current in certain applications, the techniques,as noted above, individually have a number of short comings.

What is needed is a technique for detecting a current, within, forexample, a power electronic circuit, that addresses many of theshortcomings of the prior art techniques described above. It would bedesirable for a current sensor configured according to the technique tobe capable of being reset without adversely effecting a subsequentoutput signal provided by the current sensor.

SUMMARY

In one embodiment a current sensor includes an integrated circuit dieincluding a first and second coil formed in the integrated circuit die.An integrated circuit die package holds the integrated circuit die and aconductor located within the integrated circuit die package and outsideof the integrated circuit die. The conductor receives a current from aninput terminal of the integrated circuit die package and supplies thecurrent to an output terminal of the integrated circuit die package. Theconductor and first and second coils are inductively coupled.

In another embodiment a current sensor includes a first and secondinductor in an integrated circuit die inductively coupled with aconductor outside the integrated circuit die and located in a packageholding the integrated circuit die. An integrator circuit that iscoupled to the first and second inductors is configured to generate asensed voltage responsive to a current in the conductor sensed by thefirst and second inductors.

In another embodiment a method is provided that includes receiving afirst current in a conductor located within an integrated circuit diepackage and outside of an integrated circuit die. As a result of thefirst current a signal is induced in a first and second coil in theintegrated circuit die. The induced signal is integrated in anintegrator circuit in the integrated circuit die and the integratorcircuit supplies an integrator output voltage from the integratorcircuit indicative of the first current.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention may be better understood, and its numerousobjects, features, and advantages made apparent to those skilled in theart by referencing the accompanying drawings.

FIG. 1 illustrates a prior art current sensor.

FIG. 2 illustrates a further prior art current sensor.

FIG. 3 illustrates yet another prior art current sensor.

FIG. 4 illustrates a further prior art current sensor.

FIG. 5 a illustrates a coil in close proximity with a large currentcarrying wire, according to an embodiment of the present invention.

FIG. 5 b illustrates a perspective cut away view of an integratedcircuit, including a coupled coil and wire.

FIG. 6 is a cross-sectional view of a first embodiment of an integratedcurrent sensor package.

FIG. 7 is a top view of the first embodiment of the integrated currentsensor package.

FIG. 8 is a model (equivalent circuit) of the integrated current sensorillustrated in FIGS. 6 and 7.

FIG. 9 is a cross-sectional view of an alternative embodiment of theintegrated current sensor package.

FIG. 10 is a top view of the alternative embodiment of the integratedcurrent sensor package of FIG. 9.

FIG. 11 is a model (equivalent circuit) of the alternative embodiment ofthe integrated current sensor package illustrated in FIGS. 9 and 10.

FIG. 12 is a schematic diagram of an integrated current sensor.

FIG. 13 is a schematic diagram illustrating an integrated current sensorwithin a switched power supply circuit.

FIG. 14 is a timing diagram illustrating operation of the switched powersupply circuit of FIG. 13.

FIG. 15 illustrates a further circuit for controlling a reset switch ofan integrated current sensor.

FIG. 16 is a top view of a further embodiment of the integrated currentsensor package.

FIG. 17 is a cross-sectional view of the embodiment of the integratedcurrent sensor package in FIG. 16, along the line 17-17.

FIG. 18 is a further cross-sectional view of the embodiment of theintegrated current sensor package in FIG. 16, along the line 18-18.

FIG. 19 is an electrical schematic of a circuit that includes a currentsensor implemented in a buck converter application.

FIG. 19 a is a timing diagram illustrating operation of the circuit ofFIG. 19.

FIG. 20 is an electrical schematic of a circuit that includes a currentsensor with a reset circuit, configured according to one aspect of thepresent invention.

FIGS. 20 a and 20 b are timing diagrams that illustrate operation of thereset circuit of FIG. 20.

FIG. 21 a illustrates an embodiment in which two coils are used to sensethe current instead of one.

FIG. 21 b illustrates another two coil embodiment in which the currentcarrying wire implemented on the current sensor package is “U” shaped.

FIG. 22 illustrates another view of an embodiment of a dual coil currentsensor.

FIG. 23 illustrates of how the dual coils are coupled to the integratorin an exemplary embodiment.

DESCRIPTION OF THE PREFERRED EMBODIMENT(S)

Referring now to the drawings, and more particularly to FIG. 5 a, thereis illustrated a coil 502 in close proximity with a large currentcarrying wire (or conductor) 504 such that the coil 502 and currentcarrying wire 504 act as coupled inductors. The coupled inductors, alongwith on-chip electronics, which will be discussed herein below, allowfor the creation of the V_(SENSE) signal which is proportional to aninput current i_(p) in a manner that has very low loss, is very smalland is a low cost implementation. This generally provides a bettersolution than the implementations described with respect to FIGS. 1-4.The current provided through the current carrying wire 504 may be up to,for example, 10 A. The coil 502 is placed near the current carrying wire504 in order to create inductive coupling between the wire 504 and coil502. As shown, the wire 504 only overlaps only one side of the coil 502such that the windings are all going the same way and the magnetic fluxadds together. This causes an induced current in the other side of thecoil 502 that is not overlapped by the wire 504.

Referring now to FIG. 5 b, there is illustrated a perspective cut awayview of the coil 502 and wire 504 illustrated in FIG. 5 a. In thisconfiguration one of the coupled inductors is placed within a silicondioxide layer 604 on top of a die 606 of an integrated circuit chip. Thecoil 502 consists of metal runs in, for example, an M5 layer locatedwithin the silicon dioxide layer 604. The wire 504 would rest on thesilicon dioxide layer 604 in close enough proximity to the coil 502,such that current passing through the wire 504 would induce anothercurrent within the portion of the coil 502 over which the wire 504 wasnot located.

There are multiple ways for implementing the coupled inductorconfiguration within a chip package. The first of these comprises anon-chip solution with bumping copper, as illustrated in FIG. 6. Flipchip bump houses can deposit a copper wire 602 on top of a silicondioxide layer 604 of a die 606. The copper wire 602 may comprise, forexample, 15 μm of copper. As is illustrated, the coil 502 is embeddedwithin the silicon dioxide layer 604.

Referring now to FIG. 7, there is illustrated a top view of the packageconfiguration. The copper wire 602 is placed upon the silicon dioxidelayer 604 of the die 606 (not shown in FIG. 7). The coil 502 is locatedwithin the silicon dioxide layer 604 parallel to the wire 602. Bondwires 702 connect the copper wire 602 on the die 606 with externaloutputs. The bond wires 702 typically support a maximum current of 1-2amps, thus many bond wires are required to be connected to the copperwire 602 for higher currents. Additional bond wires 704 connect portionsof the die 606 to external pins 706 of the chip. Using theabove-described package configuration, a 10 A sensor may be readilyconstructed.

Referring now to FIG. 8, there is provided a model (equivalent circuit)of the inductive coil package illustrated in FIGS. 6 and 7. The coil 802on the primary side comprises a 500 pH coil. The coil 804 on thesecondary side comprises a 2 μH coil. Connected to a first side of the500 pH coil 802 is a 1.5 mOhm resistor 808 in series with a 0.5 mOhmresistor 810. Connected to one output side of the 2 μH coil 804 is a 20kOhm resistor 812. The 0.5 mOhm resistor 810 comprises the resistanceprovided by the coil 802. Since the copper wire 602 is not too thick andlies very close to the coil 502 of the chip, coupling coefficientsbetween the copper wire 602 and the coil 502 are very good, assumingthere is a distance of approximately two microns from the coil 502(e.g., formed in an M5 layer) to the copper wire 602.

Referring now to FIG. 9, there is illustrated an alternativeconfiguration wherein a package lead frame and flip chip configurationare used. A custom package lead frame may be designed as follows. Thedie 902 is placed upside down with the silicon dioxide layer 904suspended a short distance above a large copper slug 906. The copperslug 906 may have a large cross-sectional area for low loss. In thisembodiment the slug 906 has a 200×200 μm cross-section. The die 902 issuspended above the copper slug 906 on solder balls 908, which rest ontop of a lead frame 910. When heat is applied to the circuit, the solderbumps 908 reflow causing the silicon dioxide layer 904 to rest directlyupon the copper slug 906. In this design, the die chip 902 would bebumped and then flipped.

Referring now to FIG. 10, there is illustrated a top view of theembodiment of FIG. 9 with the silicon dioxide layer 904 resting on topof the copper slug 906. Bond wires 1002 may then be connected toappropriate portions of the M5 (or other metal layers) formed within thesilicon dioxide layer 904. This design has a very low resistance.

Referring now to FIG. 11, there is illustrated a circuit representationof the embodiment illustrated in FIGS. 9 and 10. In this representation,a 200×200 μm copper slug 906 is utilized that is 3 μm away from the coil502, and the primary side includes a 520 pH coil 1102 in series with a0.5 mOhm resistor 1104. The secondary side consists of 2 μH coil 1106 inseries with a 20 kOhm resistor 1108. The coupling coefficient is reduceddue to the lower current density in the slug.

With reference to FIG. 16 there is illustrated a bottom view of afurther configuration wherein a lead on chip configuration is used. Thelead frame 1602 is connected to die 1604, by bond wires 1606. The wire1608 is connected to the die 1604 by, for example, tape 1702. The wire1608, a current carrying conductor, is inductively coupled to a coil inthe die 1604.

Referring now to FIG. 17, there is illustrated a cross-sectional view ofFIG. 16, along line 17-17. The die 1604 is connected to the wire 1608,via the tape 1702, as described previously. The lead frame 1602 connectsto the die 1604, via bond wires 1606. The tape 1702 may be, for example,approximately 75 μm thick. As is shown, the structure is containedwithin a mold compound 1704. Referring now to FIG. 18, there isillustrated a cross sectional view of FIG. 16, along line 18-18.

Turning to FIG. 12, there is illustrated a schematic diagram of theelectronic circuit necessary for recreating the V_(SENSE) signal whendetecting the current i_(p) using the coupled inductors as illustratedin FIGS. 5A and 5B. The coupled inductor 1202 comprises either of theconfiguration packages described hereinabove or, alternatively, maycomprise a different undescribed configuration package that places thecoil in close proximity with the wire to inductively couple themtogether. The primary side is modeled by inductor 1204 in series withresistor 1206. The secondary side is modeled by inductor 1208 which isconnected to a resistor 1210. The resistor 1210 is then connected toground. The other side of inductor 1208, which is connected to thenegative (inverting) input of an operational amplifier 1212, outputs theinduced current I_(n). The positive (non-inverting) input of operationalamplifier 1212 is connected to ground.

The current through the secondary is dominated by the resistive loss ofresistor 1210 and is the derivative of the primary current. Anintegrator circuit 1218 is used to integrate the induced current I_(n).The integrator circuit 1218 includes the operational amplifier 1212, acapacitor 1214 (connected between the output of operational amplifier1212 and the negative input of operational amplifier 1212) and a resetswitch 1216 (connected between the output of operational amplifier 1212and the negative input of operational amplifier 1212) in parallel withthe capacitor 1214. Thus, the current I_(n) may be determined accordingto the equation:I _(n)=(L _(m) /R ₁)(di _(p) /dt)By integrating on the capacitor 1214 an output voltage V_(SENSE) isattained according to the following equation:V _(SENSE)=1/C∫I _(n) dt=(L _(m) /R ₁ C)i _(p)In this case, L_(m), the mutual inductance, is well controlled, but canvary from part to part due to assembly variations. The capacitance Cwill vary from part to part and probably can be controlled to +/−5%accuracy. The capacitor 1214 will generally not have any appreciabletemperature coefficient. R₁ is dominated by the metal resistance of thecoil and will vary from part to part and is equal to the value of theresistor 1210 and also has a large temperature coefficient.

In order to obtain overall accuracy for the capacitance C which variesfrom part to part, factory calibration using a one time programmable(OTP) memory 1220 can be used. In a preferred embodiment, a low cost32-bit OTP memory may be utilized. The OTP memory 1220 provides acontrol variable to a programmable gain amplifier 1222. The first gainstage 1223, consisting of programmable amplifier 1222, programmableresistance 1224 and the OTP memory 1220, compensates for part to partvariations of the circuit. The OTP memory 1220 is programmed at thefactory based upon measurements made there. The programmable gainamplifier 1222 has its negative input connected to the output of theoperational amplifier 1212. A programmable resistance 1224 is connectedbetween the output of the programmable amplifier 1222 and ground. Thepositive input of programmable amplifier 1222 is connected to theprogrammable resistance 1224. The value of the programmable resistance1224, and thus the gain of the first gain stage 1223, is controlled bythe values provided from the OTP memory 1220.

A second gain stage 1226 compensates for differences in the resistancecaused by temperature variations in the device. A temperature sensor1228 and an analog-to-digital converter (ADC) 1230 are used to generatea digital temperature value to compensate for the coil resistancetemperature coefficient. The temperature sensor 1228 detects thetemperature and generates an analog representation of the temperature.The ADC 1230 converts the analog signal into a digital signal. Thedigital temperature value is provided, via a control bus 1231, tocontrol logic 1232. In one embodiment the control logic 1232 may consistof a look-up table. The look-up table would include various controlvalues associated with particular temperature values. Alternativeembodiments may include a microprocessor programmed to control theoutput according to various temperature levels or other types of digitallogic. The control logic 1232 provides a control value to theprogrammable gain amplifier 1234 and programmable resistance 1236. Thenegative input of the amplifier 1234 is connected to the output ofprogrammable amplifier 1222. The programmable resistor 1236 is connectedbetween the output of programmable amplifier 1234 and ground. Thepositive input of the amplifier 1234 is connected to the programmableresistance 1236. The particular value of the programmable resistance1236, and thus the gain of the second gain stage 1226, is controlled viathe output from the control logic 1232. The output of the amplifier 1234provides the compensated V_(SENSE) signal. The code provided by thecontrol logic 1232 is updated during the phase in which the operationalamplifier 1212 is reset responsive to a reset signal applied to switch1216. The reset signal is applied while the sensed current i_(p) iszero.

The current sensor is designed to be used in, for example, a switchedpower supply. When the current i_(p) is equal to zero, a reset signalmay be applied to switch 1216 to reset the capacitor 1214, and the logicvalue applied to amplifier 1234, via control logic 1232, is updatedresponsive to the presently sensed temperature from temperature sensor1228. Referring now to FIG. 13, there is provided one example of how toapply the reset signal to a current sensor 1302 within a buck convertercircuit. In this case, the buck converter circuit control signal φ₂ isapplied to a transistor 1304 having its drain/source path connectedbetween 12 volts and node 1306. A second transistor 1308 has itsdrain/source path connected between node 1306 and node 1310. Thetransistor 1308 is controlled by a second control signal φ₁. The currentsensor 1302 is connected between node 1310 and ground to detect currenti_(p) and provide a control signal V_(SENSE). An inductor 1312 isconnected between node 1306 and node 1314. A capacitor 1316 is connectedbetween node 1314 and ground. A load 1318 is also connected between node1314 and ground. In one embodiment, the reset signal to switch 1216 ofthe current sensor 1302 may be configured to be the control signal φ₂.

As illustrated in FIG. 14, the current i_(p) is zero when signal φ₁ goeslow and when signal φ₂ goes high at, for example, time t₁. Integrator1218 is reset during phase two when signal φ₂ goes high and the currentsensor would accept signal φ₂ as an input to drive the reset signal toswitch 1216, since the current i_(p) is zero during this time. As can beseen each time the signal φ₂ goes high, the current i_(p) is zeroenabling the reset signal to be applied to the integrator circuit 1218.

Referring now to FIG. 15, there is illustrated an alternative embodimentwherein the reset signal to the reset switch 1216 is generatedresponsive to a one-shot circuit consisting of negative glitch detectcircuit 1502 and one-shot circuit 1504. When the current i_(p) goes lowas illustrated, for example, at t₁ in FIG. 14, the negative glitchdetect circuit 1502 will detect the negative edge of current i_(p). Inresponse to this detection, the negative glitch detect circuit 1502generates a pulse to the one-shot circuit 1504. The one-shot circuit1504 then generates the reset signal to the reset switch 1216 responsiveto the pulse from the negative glitch detect circuit 1502. Other methodsfor detecting when the sensed current i_(p) goes to zero may also beutilized for generating the reset signal to reset switch 1216. Theexamples illustrated in FIGS. 13-15 are merely provided as examples ofsome embodiments thereof.

With reference to FIG. 19, a relevant portion of a buck power converter1900 is depicted that includes a current sensor 1902 that senses acurrent provided by a power source +V. Referring to FIG. 19 a, a timingdiagram 1970 depicts various waveforms used in conjunction with theconverter 1900. In this application, a signal φ₁ drives switch 1904 tocause a current ‘i’ to flow (from the power source +V through aninductor 1920) and power to be provided to a load 1912, through aninductor L1 and a capacitor C1. The current ‘i’ that flows through theinductor 1920 induces a current that flows through the inductor 1922 andresistor 1924, charging capacitor 1926. When the switch 1904 isconducting, switch 1906 is in a non-conducting state. Similarly, whenthe switch 1904 is in a non-conducting state, a signal φ₂ drives switch1906 into a conducting state. The switches 1904 and 1906 may be, forexample, implemented as enhancement-mode field-effect transistors(FETs).

As is also shown, the signal φ₂ may also function as a reset signal todrive a switch 1908 which, when conducting, shorts the capacitor 1926 ofintegrator 1910 and resets the integrator 1910 of the current sensor1902. It should be appreciated that it is desirable for the reset signalto be turned off prior to a time when the current ‘i’ again flowsthrough inductor 1920 in order to not adversely affect an output signalVOUT provided by the integrator 1910. In modern power converters thisrequirement can be difficult to meet as the falling edge of the signalφ₂ and the rising edge of the signal φ₁ may slightly overlap. In thisevent, the integrator 1910 may still be in a reset state when thecurrent ‘i’ again begins flowing through the inductor 1920. As a result,the output signal VOUT (provided by the integrator 1910) will notprovide an accurate indication of the current ‘i’ that flows through theinductor 1920. This is particularly true for a current, such as thecurrent ‘i’ shown in FIG. 19 a, where the induced current isproportional to di/dt and the change in ‘i’ is most pronounced at thestart of φ₁.

Turning to FIG. 20, a reset circuit 1950, configured according to oneaspect of the present invention, is shown implemented in conjunctionwith the current sensor 1902 of FIG. 19. A control signal, as is shownin timing diagrams 1980 and 1990 of FIGS. 20 a and 20 b, respectively,corresponds to the signal φ₂ of FIG. 19 a. It should, however, beappreciated that the control signal may be a different signal having,for example, a different polarity. The reset circuit 1950 includes aone-shot multivibrator 1960 and an AND gate (or one or more logic gates)1962. As is used herein, the term “one-shot circuit” or “one-shotmultivibrator” is a device with one stable state that, responsive to aninput signal, provides an output signal for a period of time beforereturning to the stable state. A reset signal is provided by the ANDgate 1962, responsive to the control signal and a reset_one signalprovided at the output of the multivibrator 1960. A pulse width of thereset signal corresponds to a pulse width of the shortest one of thereset and reset_one signals. As such, except for the case of extremelyshort pulse widths of control signals φ₁ and φ₂, depending upon theimplementation, the reset circuit 1950 ensures that the reset of theintegrator shorter than the control signals and thus the integratorreset is released before the current rises substantially above zero,thus allowing accurate sensing the current. It should be appreciatedthat the above-described reset circuit can be readily incorporatedwithin the same integrated circuit that includes the current sensor andthe use of φ₂ (or other signal) readily allows for external timingcontrol. Thus, one of the external pins 706 (FIG. 7) can be suppliedwith an appropriate φ₁ or φ₂ control signal for use to generate thereset signal, depending upon the location of the current sensor.Exemplary placements of the current sensor for a buck converter areillustrated in FIGS. 13 and 19.

In another embodiment, two coils are used instead of one to sense thecurrent. Referring to FIG. 21 a a top view of two coils 2101 and 2103are seen in die 2105 vertically displaced from the current carrying wire2107. The coils 2101 and 2103, shown as single turns for ease ofillustration, may be multiple turn coils. In an exemplary embodiment thewidth of the current carrying wire is 0.75 mm. The use of two coilsallows significant cancellation of stray fields from external sources,and thus more accurate current sensing in certain environments. Inanother embodiment illustrated in FIG. 21 b, the current carrying wire2109 has a different configuration and is shaped as a “U”. Potentialinterferers, wires 2111 and 2115, are also illustrated. In bothembodiments, the current carrying wires 2107 and 2109 may be copper andformed as part of the lead frame package housing the die with the dualcoils. FIG. 22 illustrates another view of an embodiment of a dual coilcurrent sensor implementation.

FIG. 23 illustrates an exemplary embodiment of how the dual coils arecoupled to the integrator. For ease of illustration, the reset circuitand the calibration and temperature compensation circuits are omittedfrom the figure. As shown in FIG. 23, the direction of current in coil2301 and 2303 causes the flux direction in the two coils to be in theopposite direction (one into the page and one out of the page), thusresulting in a substantial improvement in cancellation of interferenceover a single coil embodiment in certain applications. Note also that inFIG. 23 single turn coils are illustrated. In other embodiments thecoils may be implemented with the appropriate number of turns toadequately provide the current sensing capability.

Accordingly, a reset circuit for an integrator has been described hereinthat ensures that a reset input of an integrator is released before asensed current rises to a value that would adversely affect an outputsignal provided by the integrator in response to the sensed current.

Although various embodiments have been described in detail, it should beunderstood that various changes, substitutions and alterations can bemade therein without departing from the scope of the invention asdefined by the appended claims.

1. A current sensor, comprising: a first and second inductor in anintegrated circuit die inductively coupled with a conductor outside theintegrated circuit die and located in a package holding the integratedcircuit die; an integrator circuit coupled to the first and secondinductors and configured to generate a sensed voltage corresponding to acurrent in the conductor sensed by the first and second inductors; afirst compensation circuit for compensating the sensed voltage accordingto a first control value; a second compensation circuit for compensatingthe sensed voltage responsive to a sensed temperature; and wherein thefirst compensation circuit includes, a first programmable gain stageincluding an amplifier, for compensating the sensed voltage responsiveto the first control value, the first control value configuring thefirst programmable gain stage to compensate for part to part differencesin the integrator circuit; and a memory for storing the first controlvalue.
 2. The current sensor of claim 1, further including a lead framefor supporting the integrated circuit die.
 3. The current sensor asrecited in claim 1 wherein the integrator includes an operationalamplifier having an inverting input, a non-inverting input and an outputand a capacitor having a first terminal and a second terminal, andwherein the inverting input of the operational amplifier is coupled tothe first and second inductors and the first terminal of the capacitor,and the second terminal of the capacitor is coupled to the output of theoperational amplifier, which provides the sensed voltage.
 4. The currentsensor of claim 1, wherein the integrator circuit further comprises: anoperational amplifier; and a capacitor connected between an input of theoperational amplifier and an output of the operational amplifier.
 5. Thecurrent sensor of claim 1, wherein a first end of the first inductor iscoupled to the integrator circuit and a second end of the first inductoris coupled to a first end of the second inductor and a second end of thesecond inductor is coupled to ground.
 6. The current sensor of claim 1,wherein the second compensation circuit further comprises: a secondprogrammable gain stage including a second amplifier for compensatingthe sensed voltage responsive to a second control value, the secondcontrol value configuring the second programmable gain stage tocompensate for temperature differences; a temperature sensor forgenerating a temperature signal; and control logic responsive to thetemperature signal for generating the second control value.
 7. Thecurrent sensor as recited in claim 1 further comprising a tape disposedbetween the conductor and the integrated circuit.
 8. The current sensoras recited in claim 1 wherein the conductor is configured such that theconductor does not supply current to the integrated circuit die.
 9. Thecurrent sensor as recited in claim 1 wherein the conductor is verticallydisplaced from the first and second inductors.
 10. A method comprising:receiving a first current in a conductor located within an integratedcircuit die package and outside of an integrated circuit die; inducing asignal in a first and second coil in the integrated circuit die thatcorresponds to the first current; integrating the induced signal in anintegrator circuit in the integrated circuit die; supplying anintegrator output voltage from the integrator circuit indicative of thefirst current; compensating the integrator output voltage in a firstcompensation circuit according to a control value stored in anon-volatile memory by configuring a programmable gain stage accordingto the control value, thereby compensating for part to part differencesin the integrator circuit; and compensating the integrator outputvoltage in a second compensation circuit responsive to a sensedtemperature.